A voltage regulator is a circuit that provides a precise output voltage under varying load conditions from an unknown and possibly varying input voltage. Many different types of voltage regulators have been developed, each with its own set of advantages. This particular application is directed at a particular class of voltage regulator known as inductor-based switching voltage regulators. The two most common types of inductor-based switching regulators are Boost (output voltage greater than input voltage) and Buck (output voltage less than input voltage) switching regulators. Both Boost and Buck switching regulators are very important for battery powered applications such as cellphones.
As shown in FIG. 1A, a traditional implementation for a Buck switching regulator includes a switch 102 connected between an input voltage (VP in this case) and a node 116. A switch 104 is connected between the node 116 and the ground voltage (VN). An inductor 106 is connected between the node 116 and the output node (VOUT) of the regulator. A filtering capacitor connects VOUT to the ground voltage VN. The node VOUT is also connected to a load represented by the resistor 110.
A control circuit (described below) turns switches 102 and 104 ON and OFF in a repeating pattern. Switch 102 is driven out of phase with switch 104. Thus, when switch 102 is ON switch 104 is OFF. This causes the Buck switching regulator to have two distinct operational phases. In the first phase, shown in FIG. 1B, the switch 102 is ON. During this phase, called the charging phase the inductor 106 is connected between the battery and the output node VOUT. This causes current to flow from the battery to the load. In the process energy is stored in the inductor 106 in the form of a magnetic field. In the second, or discharge phase the switch 102 is opened (see FIG. 1C). In this phase, the inductor 106 is connected in series between ground and the load. Current supplied by the inductor's collapsing magnetic field flows to the output node VOUT and the load.
As shown in FIG. 1D, a typical Boost converter includes all of the components just described. A slightly different topology is used in which the switch 102 is placed between the inductor 106 and the output node. The Boost converter uses a similar two phase pattern of switching for its two switches.
SEPIC converters are another type of inductor-based switching regulators. SEPIC converters are more fully described in a copending U.S. patent application Ser. No. 11/933,402 entitled “High Voltage SEPIC-Converter,” now U.S. Pat. No. 8,350,546. That disclosure is incorporated in this document by reference.
To maintain its output at a constant voltage, switching regulators include control circuits that modulate the duty factor of their high and low-side switches 102 and 104, respectively. As shown in FIG. 1A, the control circuit typically includes a resistive divider formed by resistors 112 and 114 as well as an error amplifier 118, comparator 120 and break-before-make (BBM) circuit 122. The resistive divider generates a feedback voltage FB proportional to the output of the regulator. The feedback voltage FB is one of the inputs to the error amplifier 118. The second error amplifier 118 input is a reference voltage BG that is generated using any convenient technique as is well known in the relevant art. The error amplifier 118 compares the feedback voltage FB to the reference voltage BG and multiplies the difference by a gain factor to generate an output voltage EAOUT.
The error amplifier 118 output EAOUT is one of the inputs to the comparator 120. The second input to the comparator 120 is a periodic ramp voltage RAMP. The output of the comparator 120 (i.e., the comparison between the ramp voltage RAMP and the output of the error amplifier EAOUT) is a periodic square wave signal CLKV. The square wave signal CLKV is passed to the BBM circuit 122. The BBM circuit 122 generates a signal based on CLKV to drive the high-side switch 102 and a complementary signal to drive the low-side switch 104. In general, it takes a finite amount of time to turn the high and low-side switches 102 and 104, respectively, ON and OFF. For this reason, the act of turning a switch OFF is always done slightly in advance of the act of turning the other switch ON. This technique, known as break-before-make avoids the situation where both switches are ON at the same time and power is connected through the high and low-side switches to ground (a condition known as shoot through).
FIG. 1E shows the ramp voltage RAMP along with the error amplifier 118 output EAOUT. The corresponding comparator 120 output CLKV is also shown. As may be appreciated, the duty cycle of CLKV is defined by the intersection of RAMP and EAOUT. FIG. 1E also shows a higher error amplifier 118 output (labeled EAOUT′) and the effect that it has on the duty cycle of the periodic square wave signal CLKV. This is the basic feedback mechanism for the Buck regulator of FIG. 1A: decreases in the output voltage cause the feedback voltage FB to fall. This causes the error amplifier 118 output EAOUT to increase. The increase in EAOUT causes CLKV to have an increased duty cycle. This increases the duty cycle of the high-side switch 102 and decreases the duty cycle of the low-side switch 104. Thus, if the output voltage increases or decreases, the duty cycle of the high and low-side switches are adjusted in a way that compensates for the increased or decreased output.
The control loop just described is an example of what is generally referred to as voltage mode control (i.e., regulator output is regulated as a function of output voltage). In this control loop, the gain of the error amplifier determines the accuracy of regulation. A high gain amplifier keeps the deviations of the output voltage relatively small and close to ideal. A lower gain amplifier allows larger deviations to occur.
The control loop must maintain stability, that is to say, must not oscillate which would cause the output voltage to oscillate. Feedback theory provides criteria for this stability. If the gain of the control loop is plotted as a function of frequency, an element of the control loop must reduce the gain below one at some frequency. This frequency is called the gain-bandwidth (GBW) product or unity gain frequency.
A large GBW product control loop indicates that the control loop is fast and can respond to fast transients. For example, in modern microprocessors, the processor can turn on rapidly so that the supply current takes a large fast step in times approaching the switching speed of the microprocessor. A large GBW product allows the voltage regulator to respond quickly to such changes. (If the circuit does not have a large GBW product, then large output capacitors are needed to sustain the output voltage until the loop responds).
Control theory says that the phase shift around the control loop must not be greater than 180 degrees at the unity gain frequency. In fact, the circuit is not really useable if the phase shift of the control loop is near 180 degrees. It is preferable to be near 90 degrees, but in many cases 140 to 130 degrees of phase shift is acceptable.
In a voltage mode converter, the inductor-capacitor pair introduce a 180 degree phase shift by themselves at their resonant frequency: ½π*(L*C)1/2. As a result, any control loop must take this into account by removing about 90 degrees of phase shift starting at the resonant frequency.
In the parlance of control loop theory, the removal of 90 degrees of phase shift is accomplished by adding a “zero” to the control loop. If 90 degrees of phase shift is added, a “pole” is added to the control loop. The LC filter of the buck converter adds a “double pole” at the resonant frequency, to get the 180 degree phase shift.
If nothing were done except adding a wide band amplifier for control, the voltage mode converter would be unstable because of the double pole adding 180 degrees of phase shift at the unity gain frequency. For good compensation a zero must be added at the resonant frequency of the output filter to add back 90 degrees of phase shift.
In the prior art, voltage mode compensation has been generally accomplished three ways as shown in FIG. 2. The first way is placing capacitor 202 in the feedback loop. This adds a zero and a pole which are generally too close together in frequency for most cases we want to consider. This makes this technique helpful but not very useful.
Another prior art is using the parasitic resistance 204 of the filter capacitor 108 as the zero forming element. For a 20 uf filter capacitor 108, and a 30 kHz zero, this yields a parasitic resistance 204 value of 0.26 ohms which is large (for most cases) and may produce large ripple. To get to a reasonable ESR, large values of capacitance must be used, but still the ripple is a problem. Generally tantalum or other electrolytic capacitors are needed for this type of compensation. Ceramic capacitors, in general, have too low an ESR to be effective. Tantalum capacitors are generally more expensive than ceramic.
In FIG. 2 a box is shown connecting the error amplifier 118 output to the feedback node. This feedback network might be used to create a three pole, two zero circuit which can be effective to stabilize the voltage mode circuit. The resultant gain transfer curve is shown in FIG. 3. A dominant pole is introduced at about 30 Hz. At about 20 kHz a double zero is introduced, just below the resonance of the LC circuit. At about the switching frequency of the regulator, about 1 MHz, another double pole is introduced which rolls the gain off to the unity gain point above 10 MHz.
The error amplifier 118 output and FB nodes are brought to an external compensation network 206 where a dominant pole and two zeros are introduced. In order to make the system stable, a double pole must be introduced at a high frequency to roll the gain off to make the system stable. In this example, the gain bandwidth product is near 50 MHz for the amplifier. It can also be seen that the second zero's effectiveness is less than a decade. If the GBW product of the amplifier is reduced, the whole curve must be shifted to a lower frequency, which makes the regulator slower and uses larger external components.
A third compensation scheme places a low pole in the compensation network 206 such that the unity gain is reached well before the double pole of the output filter. This makes a very slow control loop.